Half bridge resonant converters, circuits using them, and corresponding control methods

ABSTRACT

A half bridge resonant converter comprises a half bridge inverter having a high side switch and a low side switch with an output defined from a node between the high side switch and the low side switch. The output connects to a resonant circuit. There are separate control circuits for generating the gate drive signals for controlling the switching of the high side switch and low side switch, in dependence on an electrical feedback parameter, each with different reference voltage supplies.

CROSS-REFERENCE TO PRIOR APPLICATIONS

This application is the U.S. National Phase application under 35 U.S.C.§ 371 of International Application No. PCT/EP2017/058660, filed on Apr.11, 2017, which claims the benefit of European Patent Application No.16165365.4, filed on Apr. 14, 2016. These applications are herebyincorporated by reference herein.

FIELD OF THE INVENTION

This invention relates to the use of half bridge resonant converters. Byway of example, such resonant converters may be used to form part of apower converter to provide AC/DC conversion, to provide DC/DCconversion, to provide AC/DC conversion with power factor correction, orto provide DC/AC conversion, i.e. inversion.

BACKGROUND OF THE INVENTION

So-called resonant converters have a resonant circuit, which can be aseries or parallel or series-parallel resonant circuit. When configuringconverters, one aim is to keep losses low. For example, resonantconverters which comprise an LLC series-parallel resonant circuit havingtwo inductances and one capacitance are well-known. Such converters havethe advantage that energy-efficient operation with relatively lowswitching losses is possible.

Resonant LLC converters are well known for use within LED drivers. Theconverters can be configured or operated as a constant current source ora constant voltage source. A constant current source can be used todrive an LED arrangement directly, thus enabling a single stage driver.Constant voltage sources can be used, for example, for LED modules whichhave further driver electronics in order to ensure a corresponding powersupply to the LEDs with a predetermined current from the output voltageprovided by the constant voltage source.

The LLC converter comprises a switching arrangement (which together withthe gate driving arrangement is generally referred to as the inverter)for controlling the conversion operation, and the switching iscontrolled using feedback or feedforward control, in order to generatethe required output.

Another function implemented within a power converter which is suppliedwith mains (or other AC) power is power factor correction (PFC). Thepower factor of an AC electrical power system is defined as the ratio ofthe real power flowing to the load to the apparent power in the circuit.A power factor of less than one means that the voltage and currentwaveforms are not in phase, reducing the instantaneous product of thetwo waveforms. The real power is the capacity of the circuit forperforming work in a particular time. The apparent power is the productof the current and voltage of the circuit. Due to energy stored in theload and returned to the source, or due to a non-linear load thatdistorts the wave shape of the current drawn from the source, theapparent power will be greater than the real power.

If a power supply is operating at a low power factor, a load will drawmore current for the same amount of useful power transferred than for ahigher power factor.

The power factor can be increased using power factor correction. Forlinear loads, this may involve the use of a passive network ofcapacitors or inductors. Non-linear loads typically require active powerfactor correction to counteract the distortion and raise the powerfactor. The power factor correction brings the power factor of the ACpower circuit closer to 1 by supplying reactive power of opposite sign,adding capacitors or inductors that act to cancel the inductive orcapacitive effects of the load.

Active PFC makes use of power electronics to change the waveform of thecurrent drawn by a load to improve the power factor. Active PFC circuitsmay for example be based on buck, boost or buck-boost switched modeconverter topologies. Active power factor correction can be single-stageor multi-stage.

In the case of a switched mode power supply, a PFC boost converter isfor example inserted between the bridge rectifier and the mains storagecapacitor. The boost converter attempts to maintain a constant DC busvoltage on its output while drawing a current that is always in phasewith and at the same frequency as the line voltage. Anotherswitched-mode converter inside the power supply produces the desiredoutput voltage or current from the DC bus.

Due to their very wide input voltage range, many power supplies withactive PFC can automatically adjust to operate on AC power for examplefrom about 110 V to 277V.

Power factor correction may be implemented in a dedicated power factorcorrection circuit (called a pre-regulator), for example placed betweenthe (mains) power supply and the switch mode power converter which thendrives the load. This forms a dual stage system, and this is the typicalconfiguration for high power LED applications (for example more than 25W). The power factor correction may instead be integrated into theswitch mode power converter, which then forms a single stage system.

In this case, there is a single resonant tank and switching arrangement,which then implements both power factor correction as well as control ofthe conversion ratio between the input and output in order to maintainthe desired output (current in the case of an LED driver) delivered tothe load.

LLC DC/DC converters are either operated at a DC supply voltage (e.g.48V in telecommunications or data center applications), or they are usedas the second stage of a mains power supply or two stage LED driver, inwhich the front end stage (the power factor correction pre-regulator)provides the power factor correction and also generates a stabilized busvoltage that forms the DC input voltage for the LLC.

An example of a resonant AC/DC converter is shown in FIG. 1.

The circuit comprises a DC input terminal 2 (labeled B in FIG. 1 and allother figures) which connects to a half-bridge having a first powerswitch 28 and a second power switch 30. The first switch and the secondswitch can be identical, and the half-bridge may for example operated ata symmetrical 50% duty cycle. These switches can be in the form offield-effect transistors.

A resonant tank circuit 25 is connected to a switch node, labeled X inFIG. 1 and all other figures between the two switches 28, 30.

Each switch has its timing of operation controlled by its gate voltage.For this purpose, there is a control block 31 (including a low voltagesupply). The block 31 receives a control signal CTRL for controlling thegate voltages and a supply voltage SUP. Feedback (not shown) is used todetermine the timing of the control of the switches 28, 30. The outputof the resonant tank circuit 25 connects to a rectifier 32 and then tothe load, in parallel with a smoothing capacitor C_(DC).

During operation of the converter, the controller 31 controls theswitches, at a particular frequency and in complementary manner.

FIG. 2 shows one more detailed example of the circuit of FIG. 1.

In this example, the resonant tank 25 is in the form of an LLC resonantcircuit, and it may be used to form a PFC stage. The circuit may thus beused as a PFC pre-regulator by having a controlled output voltage. Itcould also be used as a single stage LED driver by having a controlledoutput current.

The circuit comprises a mains input 10 which is followed by a rectifierbridge 12 having a high frequency filter capacitor 14 at the output.This generates the supply for the input terminal 2 (node B) of FIG. 1.

This example shows a converter with an isolated output. For thispurpose, the converter comprises a primary-side circuit 16 and asecondary side 18. There is electrical isolation between theprimary-side circuit 16 and the secondary side 18. A transformercomprising a primary coil 20 and a secondary coil 22 is provided for theisolation. The transformer has a magnetizing inductance 20 which alsoacts as one of the inductances of the series LLC resonant circuit. TheLLC resonant circuit 25 has a second inductance 24, and a capacitance(formed as two capacitors 26 and 27 in this example).

In an LLC circuit, the inductances and capacitor may be in any seriesorder. The inductor may comprise discrete components or it may beimplemented as leakage inductances of the transformer.

The primary-side circuit 16 comprises the half-bridge 28, 30 and theresonant tank circuit 25.

The control block 31 is shown schematically as including two voltagesources.

The secondary side 18 has the rectifier 32 which is connected downstreamof the secondary coil 22 and which can be formed, for example, by afirst diode arrangement of diodes 32 a and 32 b and a second diodearrangement of diodes 34 a and 34 b.

FIG. 2 shows a full-bridge rectifier and a single secondary coil whichcouples at its ends to the rectifier circuit. The low frequency (e.g.100 Hz) storage capacitor C_(DC) is connected between the outputs of therectifier. The LED load or other output stage is represented in thisfigure by a resistor. It comprises an LED or a plurality of LEDs.

The circuit shown in FIG. 2 is thus an AC/DC PFC converter, comprisingan AC input 10, a rectifier 12, a half bridge inverter comprising a highside switch (the first power switch 28) and a low side switch (thesecond power switch 30), wherein an output is defined from a switch nodeX between the switches. The self-oscillating LLC circuit 20,24,26,27 iscoupled to the output.

FIG. 3 shows an alternative LLC half bridge topology, as a modificationto FIG. 2 (and showing DC/DC conversion) in which the secondary coil 22has a center tap and the full wave rectifier 32 is then implemented bytwo diodes. The LLC capacitor is also shown as a single component 35.

The half bridge converter shown above may be used in a AC/DC (singlestage) PFC converter, or in a DC/DC converter, or in an AC/DC converterwithout implementing power factor correction. In the case of a DC/DCconverter, the rectifier bridge 12 and filter capacitor 14 are simplyomitted as in FIGS. 1 and 3. The half bridge converter may also be usedin a DC/AC converter, i.e. a resonant half bridge inverter. The resonanttank circuit 25 may also be of other types, and the invention is notlimited to LLC circuits.

In the case of DC/AC conversion, a load is connected to the output ofthe resonant tank circuit whereas in case of DC/DC or AC/DC conversionthe load is connected via the active or passive rectifier network to theresonant tank circuit.

Half bridge resonant converters are used already in many applicationslike DC/AC converters for lighting applications, e.g. low- andhigh-pressure discharge lamp circuits, and DC/DC converters, e.g. DCpower supplies and LED drivers.

The control block 31 drives the two power switches 28, 30 to conduct inan alternating sequence on and off, with a small non-conduction phase(dead time) used to avoid cross conduction of the power switches. A highgate drive signal turns on one switch and turns off the other switch anda low gate drive signal turns off the one switch and turns on the otherswitch. The advantage of using a resonant half bridge converter is thatthe current flowing into the switch node X has a phase lag, with respectto the switch node voltage Vx, and can serve to discharge the(parasitic) output capacitance of the switch before it will beswitched-on.

This method is referred to as Zero Voltage Switching (ZVS) and implieszero switching losses due to the parasitic output capacitance. If theoutput current is not large enough or even zero and further depending onthe operation conditions (in terms of the half bridge, output, andresonant capacitor voltage), discharging of the parasitic outputcapacitance will be partly or even completely achieved by the powerswitch which results in hard switching. This results in switching losseswhich depend on the switching frequency, the parasitic outputcapacitance of the switch and the voltage across the parasiticcapacitance at switch-on. In order to reduce the switching losses,Valley Switching (VS) can be applied which causes a switch to switch-onat the minimum voltage across it. Valley switching can be implemented bymeans of an end-of-slope detection mechanism. Zero voltage switching isa special case of valley switching where the voltage is minimal andzero.

In order to avoid critical timing of the switch turn-on, a diode can beplaced in anti-parallel to the power switch 28, 30 if a bipolar junctiontransistor is used. This anti-parallel diode may be omitted for a MOSFETbecause it already has a body diode inside. The anti-parallel diode willstart conducting if the switch is not switched on immediately afterdischarging of the voltage across the switch has occurred, and then theswitch can take over a bit later when it is eventually turned on.

Zero voltage switching ensures that the voltage across a switch is zerobefore it will be switched on and as such eliminates switching losseswhich makes high frequency (HF) operation possible. HF operation enablesa reduction in the size of capacitive and inductive components used inthe resonant tank circuit which makes smaller and cheaper designspossible.

In these circuits, the first power switch 28 connected to the rectifiedmains (or other DC input) needs a drive signal which should be close tothe switch node voltage Vx which can range from ground up to the highrectified mains voltage (or other DC voltage) at terminal 2 forswitching on and off. This means that a level shifter function isneeded.

FIG. 4 shows a driver transformer for this purpose. There are twosecondary coils 40, 42 each connected across the source and drain of arespective one of the power switches 28, 30. The secondary coil 40 setsthe gate voltage of the first power switch 28 relative to the switchnode X and the secondary coil 42 sets the gate voltage of the secondpower switch 30 relative to ground. The secondary coils have oppositepolarity to provide the complementary switching.

FIG. 5 shows a high voltage level shifting integrated circuit 50 havinga level shifting unit 52 and gate driver circuits 54, 56 for the firstand second power switches 28, 30.

By way of example, it may be desired to implement switching frequenciesas high or even higher than 1 MHz and with a maximum rectified mainsvoltage of 375V. This voltage level should be able to be raised to atleast 500V whilst still preventing damage of the switches and drivecircuits during mains surges.

The two level shift implementations shown have drawbacks.

A transformer level shifter can be used for both low frequency and highfrequency operation and an isolation voltage of 500V can indeed beachieved. However, it draws four times more power than needed to supplythe gate charge of the power switch and the unavoidable leakageinductance in the transformer causes ringing. In the case of lowfrequency applications, the extra dissipation might be not a problem butfor high frequency applications, the additional power dissipation willbe an issue. Additionally, the ringing suppression measures which may berequired cause severe turn on/off delays which might be not acceptablein high frequency operations.

The high voltage IC level shifter is currently only available for lowfrequency operation, not higher than about 1 MHz.

This invention relates to an improvement to the system for generatingand applying the control signals to the power switches of the halfbridge converter to address the issues explained above.

SUMMARY OF THE INVENTION

The invention is defined by the claims.

Examples in accordance with a first aspect of the invention provide ahalf bridge resonant converter, comprising:

a pair of DC voltage lines arranged to provide a bus voltage, whereinthe pair of DC voltage lines comprises a high voltage line and a lowvoltage line;

a half bridge inverter in series between the high voltage line and thelow voltage line, wherein the half bridge inverter comprises a high sideswitch and a low side switch, wherein an output of the half bridgeinverter is defined from a node between the high side switch and the lowside switch;

a resonant circuit coupled to the output of the half bridge inverter;

a first control circuit for generating a gate drive signal forcontrolling the switching of the high side switch, wherein the firstcontrol circuit is arranged to control a duty cycle of the high sideswitch by increasing the on-time of the high side switch if an averageswitch node voltage is lower than a fraction of the bus voltage and bydecreasing the on-time of the high side switch if the average switchnode voltage is higher than the fraction of the bus voltage; and

a second control circuit for generating a gate drive signal forcontrolling the switching of the low side switch in dependence on anelectrical feedback parameter.

The first and second control circuits may be considered to be part ofthe inverter.

This converter makes use of separate circuits for generating the gatedrive signals for the two power switches of the inverter, each withtheir own voltage domain. In this way, the circuits can use primarilylow voltage components, with the number of high voltage componentsreduced to a minimum.

One circuit is referenced to ground and the other is referenced to theswitch node between high side switch and the low side switch. Thiseliminates the need for a driver transformer or high voltage integratedcircuit. The two control circuits may be designed to provide zerovoltage switching when possible or valley switching if not, in order toeliminate or reduce switching losses whenever possible.

The converter may further comprise the first control circuit, arrangedto turn on the high side switch after the low side switch is turned offand a dead-time has elapsed.

The turn on of the high side switch is set after the low side switch hasturned off and a dead-time has elapsed. This allows for an easyswitching control implementation.

The converter may further comprise the second control circuit, arrangedto turn on the low side switch after the high side switch is turned offand a dead-time has elapsed.

The turn on of the low side switch is set after the high side switch hasturned off and a dead-time has elapsed. This allows for an easyswitching control implementation.

The converter may further comprise the second control circuit, arrangedto control an output power and/or a power factor of the half bridgeresonant converter by controlling the low side switch.

The control topology as proposed in the application is specificallysuited for an easy implementation for controlling the output powerand/or providing a high power factor.

The converter preferably further comprises a first generating circuitfor generating the first supply voltage from the high voltage line andfrom the voltage at the switch node between the high side switch and thelow side switch, and a second generating circuit for generating thesecond supply voltage from the low voltage line and from the voltage atthe switch node between the high side switch and the low side switch.

In this way, the high supply voltage for each control circuit is derivedfrom the two supply voltages and the voltage at the switch node betweenthe high side switch and the low side switch. The generating circuitsmay use high voltage components, but then the supply voltages generatedenable the control circuits to be formed as low voltage circuits.

The generating circuit may be used only during startup of the circuit,before oscillation has settled. Once the circuit is oscillating part ofthe generating circuits may be disabled.

The first generating circuit may comprise:

a first input for receiving a voltage between (i) the switch nodebetween the high side switch and the low side switch and (ii) theresonant circuit;

a charge pump circuit for converting the AC voltage at the first inputinto a DC voltage and storing it on a first output capacitor as theoutput of the first generating circuit at the first supply voltage; and

a supply transistor between the high voltage line and the output of thefirst generating circuit.

The supply transistor may be the only high voltage component required.It is used to provide a power supply during startup. The feedbackvoltage from the resonant circuit, i.e. the first input, can then beused to provide the power supply for controlling the half bridgeswitching.

The second generating circuit may comprise:

a second input for receiving a voltage between the resonant circuit andthe low voltage line;

a charge pump circuit for converting the AC voltage at the second inputinto a DC voltage and storing it on a second output capacitor as theoutput of the second generating circuit at the second supply voltage;and

a supply transistor between (i) the switch node between the high sideswitch and the low side switch and (ii) the output of the secondgenerating circuit.

Again, the supply transistor may be the only high voltage componentrequired and is used to provide a power supply during startup. Thefeedback voltage from the resonant circuit, i.e. the first input, canthen be used to provide the power supply for controlling the half bridgeswitching.

The first and second generating circuits may employ dedicated auxiliarywindings added to the transformer (when one is used). These can beconsidered as floating high frequency AC supply voltages connected witha first terminal to either ground (for the first generating circuit) orthe switch node (for the second generating circuit) while the respectivesecond terminal is connected to a rectifier diode supplying the voltagedomain.

The first control circuit may comprise:

a first end of slope detection circuit having as input the high voltageline;

a first latch element triggered by the end of slope detection circuitand which generates a first control signal for switching the high sideswitch to a first state; and

a first signal generator for generating a second control signal forswitching the high side switch to a second state.

In this circuit, the first state may be an ON state. The ON transitionis thus generated by end of slope detection, which ensures that theswitch will be turned on at the minimum voltage across its parasiticoutput capacitance. This enables ZVS or VS to be implemented.

The first signal generator may have a reference input for controllingthe duration of the first state. This reference input may be generatedby a resistive divide between the high and low voltage lines. Itcontrols the ON time of the high side switch.

The second control circuit may comprise:

a second end of slope detection circuit having as input the switch node;

a second latch element triggered by the end of slope detection circuitand which generates a third control signal for switching the low sideswitch to a first state; and

a second signal generator for generating a fourth control signal forswitching the low side switch to a second state.

Again, the first state may be the ON state and the second state is thenthe OFF state. The second signal generator for example has a feedbackcontrol input for controlling the duration of the first state independence on the electrical feedback parameter.

The electrical feedback parameter for example comprises a voltage whichis dependent on the output current delivered by the converter to a load.

A transformer may be provided between the resonant circuit and an outputload. This enables isolation of the output. The resonant circuit forexample comprises an LLC circuit.

The invention also provides an apparatus comprising:

the converter as defined above; and

the output load.

The output load may be an LED arrangement of one or more LEDs.

Examples in accordance with another aspect of the invention provide aconversion method, comprising:

operating a half bridge inverter comprising a high side switch and a lowside switch between a DC high voltage line and a DC low voltage line,using a gate drive signal and providing an output from a switch nodebetween the high side switch and the low side switch;

providing the output of the half bridge inverter to a resonant circuit;

generating a gate drive signal using a first control circuit, forcontrolling the switching of the high side switch in dependence on anelectrical feedback parameter, wherein the first control circuit has asits reference voltage supply the voltage at the switch node between thehigh side switch and the low side switch and a first supply voltagegreater than the voltage at the switch node between the high side switchand the low side switch; and

generating a gate drive signal using a second control circuit, forcontrolling the switching of the low side switch in dependence on theelectrical feedback parameter, wherein the second control circuit has asits reference voltage supply the low voltage line and a second supplyvoltage greater than the voltage at the low voltage line.

This method makes use of separate circuits for generating the gate drivesignals for the two power switches of the inverter, each with their ownvoltage domain. In this way, the circuits can use primarily low voltagecomponents, with the number of high voltage components reduced to aminimum.

Another example of a method with another aspect of the invention providea conversion method, comprising: operating a half bridge invertercomprising a high side switch and a low side switch between a DC highvoltage line and a DC low voltage line providing a bus voltage, using agate drive signal and providing an output from a switch node between thehigh side switch and the low side switch;

providing the output of the half bridge inverter to a resonant circuit;

generating a gate drive signal using a first control circuit, forcontrolling a duty-cycle of the high side switch by increasing theon-time of the high side switch if an average switch node voltage islower than a fraction of the bus voltage and by decreasing the on-timeof the high side switch if the average switch node voltage is higherthan the fraction of the bus voltage; and

generating a gate drive signal using a second control circuit, forcontrolling the switching of the low side switch in dependence on anelectrical feedback parameter.

The method may further comprise generating the first supply voltage fromthe high voltage line and from the voltage at switch node between theswitches, and generating the second supply voltage from the low voltageline and from the voltage at switch node between the switches.

BRIEF DESCRIPTION OF THE DRAWINGS

Examples of the invention will now be described in detail with referenceto the accompanying drawings, in which:

FIG. 1 shows the general architecture of a half bridge resonantconverter;

FIG. 2 shows one more specific example of a half bridge resonantconverter used in a resonant AC/DC converter which forms a PFC stage;

FIG. 3 shows another more specific example of a half bridge resonantconverter used in a resonant DC/DC converter;

FIG. 4 shows a first known level shifting arrangement for generatinggate drive signals;

FIG. 5 shows a second known level shifting arrangement for generatinggate drive signals;

FIG. 6 shows an example of a circuit in accordance with the invention,in schematic form;

FIG. 7 shows an example of a circuit in accordance with the invention,in more detail;

FIG. 8 shows an example of the implementation of the high side controlcircuit;

FIG. 9 shows a timing diagram for the operation of the circuit of FIG.8;

FIG. 10 shows an example of the implementation of the low side controlcircuit;

FIG. 11 shows a timing diagram for the operation of the circuit of FIG.10;

FIG. 12 shows an example of the implementation of the high side supplygeneration circuit; and

FIG. 13 shows an example of the implementation of the low side supplygeneration circuit.

FIG. 14 shows another example of AC/DC LLC converter circuit which mayuse a converter of the invention; and

FIG. 15 shows the controller in FIG. 14 in more detail for a singlethreshold voltage implementation.

DETAILED DESCRIPTION OF THE EMBODIMENTS

The invention provides a half bridge resonant converter comprising ahalf bridge inverter having a high side switch and a low side switchwith an output defined from a switch node between the high side switchand the low side switch. The output connects to a resonant circuit.There are separate control circuits for generating the gate drivesignals for controlling the switching of the high side switch and lowside switch, in dependence on an electrical feedback parameter, eachwith different reference voltage supplies.

FIG. 6 shows converter using a half bridge topology with an LLC resonanttank circuit 25 and a full wave rectifier 32 controlled by two localcontrol circuits.

The converter is supplied by a pair of DC voltage lines comprising a DChigh voltage line 60 (node B) and a low voltage line 62, e.g. ground. Asin the examples above, the half bridge inverter comprises a high sideswitch 28 and a low side switch 30 in series between the high voltageline 60 and the low voltage line 62. The output of the half bridgeinverter is defined from the switch node X between the high side switchand the low side switch.

A first control circuit 64 generates a gate drive signal for controllingthe switching of the high side switch 28 in dependence on an electricalfeedback parameter (as discussed below). The first control circuit 64has as its reference voltage supply the voltage at the switch node X anda first supply voltage 65 greater than the voltage at the switch node X.As explained below, the first supply voltage is generated from the mainpower supply before the circuit is oscillating but it is generated byfeedback from the resonant circuit during oscillation, thereby savingpower.

A second control circuit 66 generates a gate drive signal forcontrolling the switching of the low side switch 30, again in dependenceon the electrical feedback parameter, wherein the second control circuit66 has as its reference voltage supply the low voltage line 62 and asecond supply voltage 67 greater than the voltage at the low voltageline. Again, the second supply voltage is generated from the main powersupply before the circuit is oscillating but it is generated by feedbackfrom the resonant circuit during oscillation, thereby saving power.

The feedback may directly control the timing of only one of theswitches. However, it will then indirectly control the other in thatthere is a switching sequence between the two switches. Thus, theoverall control block 31 may be considered to be the combination of thecontrol circuits 64 and 66, and feedback control (shown as input FB) isused by the controller. The switching frequency is typically controlled,either based on a frequency control circuit or based on thresholddetection of a self-oscillating resonant tank circuit 25.

This arrangement avoids the need for a level shifter transformer andalso enables high frequency operation by using separate low voltagecircuitry locally connected to both switches.

FIG. 7 shows an implementation of the circuit in more detail.

A first generating circuit 70 is used for generating the first supplyvoltage 65 from the high voltage line 60 and from the voltage at theswitch node X. A second generating circuit 72 is used for generating thesecond supply voltage 67 from the low voltage line 62 and from thevoltage at the switch node X.

The first generating circuit 70 has a first input 71 for receiving avoltage SUP_(HS) between the switch node X (Vx) and the resonantcircuit. As shown, this high side supply voltage SUP_(HS) is derivedfrom a switch node between a series output capacitor 74 and the resonantcircuit.

Capacitors 74 and 76 function as a capacitive voltage divider withrespect to the resonant capacitor Cs. If for example the peak to peakvoltage across Cs is 500V and Cs is 1 nF, the capacitor 74 may be about20 nF to achieve a maximum voltage drop of about 25V for the supply,which practically turns into a lower value at the first supply voltage65 (LV_(HS)) depending on the charge pump impedances (the charge pump isexplained below with reference to FIG. 12) and the load. The voltageSUP_(HS) is an AC voltage with respect to the switch node X.

This voltage is used to generate the power supply for controlling theswitching of the high side switch, once the circuit is oscillating.

The second generating circuit 72 has a second input 73 for receiving alow side supply voltage SUP_(LS) between the resonant circuit and thelow voltage line 62. In particular this low side supply voltage SUP_(LS)is derived from a switch node between the resonant circuit and a lowside series capacitor 76, which capacitor then connects to the lowvoltage line 62.

This voltage is used to generate the power supply for controlling theswitching of the low side switch, once the circuit is oscillating.

This arrangement achieves valley switching by using an end-of-slopetrigger action as explained in more detail below. This is implementedusing a capacitor 78 (C_(ONHS)) between the high voltage line 60 and thefirst control circuit 64, and a capacitor 79 (C_(ONLS)) between theswitch node X and the low side control circuit 66.

The high side control circuit 64 receives power from the resonant tankcircuit by means of the capacitor 74 and from the first generatingcircuit 70, and the low side control circuit 66 receives power from theresonant tank circuit by means of the capacitor 74 and the secondgenerating circuit 72. Both local supplies need to be powered byalternative means before the start-of-oscillation occurs, and thisrequires a high voltage transistor for each. These high voltagetransistors reside inside the generating circuits 70, 72 as shown below.

All the circuitry described may be implemented using low voltagediscrete components, low voltage ICs or a combination of both, exceptfor the end-of-slope sense capacitors 78, 79 and the high voltage supplytransistor inside each generating circuit.

The capacitor arrangement (C_(ONHS), C_(ONLS)) is used to ensure thatthe power switch will be switched on at the minimum voltage across itsown (parasitic) output capacitance. This implies zero voltage switchingwhen there is a sufficiently large current at the moment of switch-offof the complementary power switch.

Communication between the high side and low side drive circuit isestablished using the switch node voltage (Vx) information and by meansof resistors R_(SS1) and R_(SS2). These form a potential divider betweenthe high voltage line and the low voltage line, providing a 1:2 divisionof the voltage at terminal B. The output is used as a reference withwhich the average switch node voltage is compared, and the average isthus controlled to be half of the voltage at node B. This providesbalancing control.

The on-signal is not directly transmitted but the previous off-signalcauses the switch node commutation which in turn is sensed by the othervoltage domain. Secondly, the switch node average voltage is explicitlycontrolled by one domain which implies the same on-time as the otherdomain but without direct transmission of any on-time signal between thevoltage domains.

In the example shown, the voltage generated by the resistors R_(SS1) andR_(SS2) is used to control the on-time of the high side) switch 28. Theon-time of the lower switch is then controlled by means of a feedbacksystem. A feedback voltage V_(CTRL) is compared with a reference levelV_(SET) to provide error based feedback control. In this example, thefeedback voltage V_(CTRL) is proportional to the output current I_(OUT)of the converter. The output current I_(OUT) supplies the LED string,connected between LED+ and LED−, and the output filter capacitor C_(DC)which provides 100 Hz ripple reduction.

Note that the roles of the lower switch and upper switch control may beexchanged.

The feedback voltage is the voltage across an output resistor ROUT inresponse to the output current I_(OUT) and the voltage V_(CTRL) iscontrolled to be equal to the reference V_(SET). This control loop thuscontrols the output current of the converter I_(OUT).

The advantages of using these local drive circuits are that only cheapand fast low voltage components are involved to control and drive thelocal power switch except for a small and inexpensive capacitor of a fewpF across each power switch, for example a 500V capacitor (i.e. thecapacitors 78, 79). This results in automatic ZVS (Zero VoltageSwitching) or VS (Valley Switching) for both power switches.Additionally, a simple and cheap local supplementary low voltage supplyis derived from the resonant tank circuit using low voltage components.The initial supply voltage before start-of-oscillation can then besupplied e.g. via a low cost 500V BJT (Bipolar Junction Transistor)capable of handling a few mA. A MOSFET may instead be used.

Except for a few high voltage components, the local control circuitrycan be integrated using a low voltage IC process (e.g. 10V to 25V). Thesame IC may be used twice for driving the high side and low side gate.The respective control task can be selected e.g. via externalcomponents.

FIG. 8 shows an implementation of the high side (first) control circuit64 and FIG. 9 shows the operation using waveforms.

The first control circuit comprises a first end of slope detectioncircuit 80 having as input the high voltage line (node B). A first latchelement 82, in this example in the form of a D-type flip flop, istriggered by the end of slope detection circuit 80 and generates a firstcontrol signal HS_ON for switching the high side switch to a first, ON,state. It is provided to the clock input of the flip flop.

A first signal generator 84 is used for generating a second controlsignal HS_OFF for switching the high side switch to a second state. Itsinverse is provided to the (inverse) reset input of the flip flop 82.The first signal generator 84 has a reference input from the resistivedivider R_(SS1), R_(SS2) for controlling the duration of the firststate.

In this way, the high side control circuit 64 uses an end-of-slopedetection mechanism formed by the capacitor 78 and a diode and resistorcircuit (D_(NEG), D_(POS), D_(OFF) and R_(CLK)) which triggers thepositive edge triggered flip flop 82 ON at the end of the negative slopeof V_(B,X) (i.e. V_(B) relative to V_(X)) and switches the high sidepower switch 28 on via the output from flip flop 82, GATE_HS.

The high side power switch 28 is switched off by means of the controlsignal HS_OFF in a manner which provides balancing.

FIG. 9 shows the control signals arising in the circuit.

The first positive pulse of V_(B,X) is with the high side OFF and lowside ON, so that V_(X) is pulled down by the low side switch hence V_(B)is greater than V_(X). The low side switch is turned ON only after thehigh side switch as turned OFF, as shown.

The start of the negative slope in the voltage V_(B,X) (caused byturning the low side off, LS_OFF) pulls down HS_ON (relative to thevoltage at switch node X), and the next rising edge only arises at theend of the slope. Once the high side switch is ON there is the dip tozero in the voltage V_(B,X).

FIG. 10 shows an implementation of the low side (second) control circuit66 and FIG. 11 shows the operation using waveforms.

The second control circuit comprises a second end of slope detectioncircuit 90 having as input the switch node X. A second latch element 92,again in the form of a positive edge triggered D-type flip flop, isclocked by the end of slope detection circuit 90 and it generates athird control signal LS_ON for switching the low side switch to a first,ON, state.

A second signal generator 94 is used for generating a fourth controlsignal LS_OFF for switching the low side switch to a second state. Itsinverse is provided to the (inverse) reset input of the flip flop 92.The second signal generator 94 receives the feedback control input VCFRL for controlling the duration of the first, ON, state in dependenceon the feedback.

The low side control circuit 66 thus also uses an end-of-slope detectionmechanism formed by the capacitor 79 and a diode and resistor circuit(D_(NEG), D_(POS), D_(OFF) and R_(CLK)) which triggers the positive edgetriggered flip flop ON at the end of the negative slope on the switchnode voltage V_(X) and switches the low side power switch 30 ON via theflip flop output GATE_LS.

FIG. 11 shows the control signals arising in the circuit.

The first dip in V_(X) is with the high side OFF and low side ON, sothat V_(X) is pulled down by the low side switch.

The start of the negative slope in the voltage V_(X) (caused by turningthe high side off, HS_OFF seen in FIG. 9) pulls down LS_ON, and the nextrising edge only arises at the end of the slope. Once the low sideswitch is ON there is the dip to zero in the voltage V_(X). The rise inthe voltage at switch node X is triggered by the LS_OFT signal whichturns off the low side switch, with timing based on feedback controlimplemented by the second signal generator 94.

FIG. 12 shows the first (high side) generating circuit. It comprises asupply transistor 120 between the high voltage line (node B) and theoutput LV_(HS) of the first generating circuit. A charge pump circuit122 is used for converting the AC voltage SUP_(HS) at the first inputinto a DC voltage and storing it on a first output capacitor Co as theoutput of the first generating circuit at the first supply voltage. Thelow voltage rail for the circuit is the switch node X.

The second (low side) generating circuit is the same but operates in adifferent voltage domain. It has a supply transistor 130 between theswitch node X and the output LV_(LS) of the second generating circuit.

A charge pump circuit 132 is used for converting the AC voltage at thesecond input SUP_(LS) into a DC voltage and storing it on a secondoutput capacitor Co as the output of the second generating circuit atthe second supply voltage. The low voltage rail for the circuit is thelow voltage line 62.

Thus, in both cases, the local power supply has a high voltage lowcurrent transistor (BJT or MOSFET) which charges an output capacitor Cobefore the start-of-oscillation occurs. The gate signal to thetransistor GATE_LS GATE_HS is controlled to switch off the transistorwhen oscillation starts.

In this way, the transistor may be considered to be a primary supply forstartup, and the feedback from the resonant circuit provides a secondarysupply which is used once the circuit is in oscillation.

The charge pump converts the AC peak-to-peak voltage across the inputcapacitor (C_(SL) or C_(SH)) to a DC voltage across Co. A Zenerfunction, represented by diode 124, 134, limits the output voltage incase of excessive supply.

As mentioned above, the converter may be used within an AC/DC converter,a DC/DC converter or a DC/AC converter. It may be used in a front endPFC circuit.

The front end PFC application of an LLC converter poses several problemsfor the feedback control of the inverter switch arrangement, whichcannot be mastered by the conventional frequency control approach. Thismainly has to do with the high gain ratio requirements. The gain ratiois the ratio between the maximum and the minimum gain.

The gain ratio problem can be relaxed if instead of the switchingfrequency, a threshold for an LLC state variable is used as themanipulating variable for controlling the input current. For example athreshold voltage may be set for the capacitor voltage across thecapacitor of the LLC tank. Alternatively, the transformer voltage, orthe transformer input current can also be used.

FIG. 14 shows an AC/DC LLC converter circuit using the capacitor voltageas the control variable.

As in FIG. 1, the circuit has an AC mains input 10 followed by arectifier 12. The switches 28, 30 of the half bridge inverter arecontrolled by a gate driver 140 which is controlled by a controller 142.The controller outputs a gate drive signal GS.

The controller is provided with a threshold value which in this exampleis the threshold (or reference) capacitor voltage vC_ref. The controller142 receives the measured quantity i.e., the actual resonant capacitorvoltage vC, and processes the switching scheme for the gate driver 140that in turn controls the inverter 28, 30 and the switch node voltageVx, i.e. the voltage at the output of the half bridge inverter.

The controller thus has an outer control loop 144 for setting athreshold level for the electrical feedback parameter (the capacitorvoltage) in dependence on the output voltage vo in this example and theinput voltage and current vm, im, and an inner control loop 142 forcomparing the electrical feedback parameter with the threshold to derivethe gate drive signal.

The outer control loop 144 implements output control as well asimplementing PFC, and the inner control loop 142 derives the switchingcontrol signal.

FIG. 15 shows the controller 142 in more detail. The measured capacitorvoltage vC is compared with the reference vC_ref by comparator 150, andthe comparison result is used to reset a flip flop 152 which generatesthe output for the gate driver 140. A delay element 154 provides adelayed set pulse so that the reset operation has a fixed duration(which is a function of the clocking speed of the flip flop).

This feedback system comprises a high frequency control loop implementedby the inner control loop 142.

The outer low frequency controller 144 receives the mains voltage vm,the actual mains current im and output voltage vo and its set pointvo_ref and processes, in accordance with the power factor needs, themanipulating value of vC_ref for the switching unit.

In this example, there is only one threshold value (vC_ref) that iscompared to a state variable (here vC). If the state variable exceedsthe threshold, the flip-flop 152 in the controller 142 is reset and theinverter is switched off via the gate driver, i.e., the switch nodevoltage is set to its minimum value.

The inverter is switched on again a certain time after the switch offevent. This time adapted to result in a symmetric operation i.e., at aduty cycle of the switch node of 0.5.

The capacitor voltage is one example of state variable which is used asa control input for the control of the inverter switching. Analternative state variable is the transformer voltage. The scheme issimilar but signs have to be changed. For example, if a threshold isexceeded the flip flop 152 in the controller 142 has to be switched on.

In another scheme, there are two thresholds. The inverter is switchedoff (on) once the state variable exceeds a first upper threshold and theinverter is switched on (off) if the state variable passes a secondthreshold. Here, the second threshold is a function of the firstthreshold and the input voltage.

In this way, the control circuit is adapted to set a first threshold ofthe electrical feedback parameter for turning on the gate drive signaland a second threshold of the electrical feedback parameter for turningoff the gate drive signal.

Instead of using a transformer as isolation means, isolating capacitorsmay be used as well. For example, by using an extra isolating (e.g. DCblocking) capacitor between the inverter switch node and thetransformer, and another between the other primary side winding terminaland the midpoint of the resonant capacitors.

Alternatively, in order to save components, the resonant capacitors canalso be designed for isolating from the mains voltage (Y-capacitors).Here the above mentioned state variable (vC) cannot be accessed directlyany longer but can be derived by measuring and integrating the currentinto the isolating capacitors.

In any of these configurations, the transformer need not to be isolatingand can be simplified, depending on the end use of the circuit.

There are various drive schemes that may be used for driving the highside and low side switches. Furthermore, the resonator may beself-oscillating or it may be driven by a frequency control circuit.

In general, a control scheme is required to drive the switches 28, 30into their on- and off-states such that the output voltage or current isregulated to a certain desired value or range of values and for a PFCcircuit also to implement power factor correction.

In order to exploit best the powertrain and to achieve the maximumefficiency, it is desired to operate the converter symmetrically (atleast at full load) and to load the transformer and the rectifier in thesecondary side equally. In the case of a transformer with center-tappedoutput windings that are symmetric in terms of turn-ratios and leakages,secondary side symmetry can be assured if the duty cycle of thehalf-bridge (i.e., its switch node) is kept at 50%.

There are basically four transitions that the control scheme musthandle:

1. Turn-on of the high-side MOSFET 28;

2. Turn-on of the low-side MOSFET 30;

3. Turn-off of the high-side MOSFET 28;

4. Turn-off of the low-side MOSFET 30.

There are several known schemes that may be used in order to achievethis.

A. Von-Voff is a control scheme where transition number 4 is initiatedwhen some state variable crosses a certain threshold voltage (Von).Following this, the control waits for a certain time (i.e., thedead-time) before starting transition 1. This dead-time ensures thatcross-conduction, or shoot-through, does not occur. The half-bridge isnow in the on-state. Eventually, either the same or a different statevariable will cross a second threshold (Voff), and transition number 3will be initiated. As with the transition to the half-bridge on-state,there will then be a dead-time before transition number 2 is initiated.The half-bridge is now in the off-state, and then the procedurecontinues from the beginning once more. The actual values of the twothresholds are determined by an outer control loop in order to yield thecorrect output. This is a Von-Voff scheme in that voltage thresholdcontrols the switching on and off.B. Von-Ton is a control scheme where transition number 4 is initiatedwhen some state variable crosses a certain threshold voltage (Von). Asin case A, the dead-time is allowed to pass before starting transitionnumber 1. Transition number 3 is initiated based on a certain timeinterval elapsing. This may be a fixed interval, or a controlledinterval. After the dead-time has then elapsed, transition number 2 isinitiated, and then the procedure continues from the beginning oncemore. The actual value of the voltage threshold is determined by anouter control loop in order to yield the correct output, and the timethreshold may be fixed or controlled dynamically. This is a Von-Tonscheme in that a voltage threshold controls the turning on (after a deadtime) and the time duration of the on period of the half bridge is thencontrolled.C. Voff-Toff is similar to case B, except that the voltage and timethresholds define the off and on transitions of the half-bridge,respectively. Transition number 3 is initiated when some state variablecrosses a certain threshold voltage (Voff). The dead-time is allowed topass before starting transition number 2. Transition number 4 isinitiated based on a certain time interval elapsing. After the dead-timehas then elapsed, transition number 1 is initiated, and then theprocedure continues from the beginning once more. As in case B, theactual value of the voltage threshold is determined by an outer controlloop in order to yield the correct output, and the time threshold may befixed or controlled dynamically. This is a Voff-Toff scheme in that avoltage threshold controls the turning off and the time duration of theoff period of the half bridge is controlled (i.e. between turning offthe high-side MOSFET and turning it on again after the time duration anddead-time).

In cases B and C, it is most often desirable to control the on or offtime such that it matches the off or on time respectively, i.e., it isusually beneficial to operate with a 50% duty cycle as mentioned above.There is no level shifter, gate drive transformer or any other meansthat could send synchronized signals between the first and secondvoltage domain of the first and second local control circuit. In orderto yet allow constant duty cycle operation, the first control circuit(64) controls the duty-cycle via controlling the average switch nodevoltage (Vx) to be for example a fraction, preferably half, of the busvoltage. This is achieved by increasing the on-time of the high sideswitch if the measured and filtered switch node voltage (x) is lowerthan half the bus voltage and decreasing the on-time of the high sideswitch if the measured and filtered switch node voltage (x) is higherthan half the bus voltage.

In summary, the four switching signals that have to be generated percycle can be divided into two groups: The two turn-on signals may beconsidered as “slave” signals that are generated in response to the two“master” (i.e., turn-off) signals. The turn-on of the high side switchfollows the turn-off of the low side switch after a certain dead timeand the turn-on of the low side switch follows the turn-off of the highside switch after a certain dead time. The synchronization is achievedby observing the switch node voltage transition by means of theend-of-slope-detection circuits. In contrast, the “master” signal of thefirst control circuit for controlling the switching of the high sideswitch is generated based on the filtered (average) switch node voltage(Vx). Thus, Vx is used in two manners in order to synchronize theswitching of the two voltage domains; in terms of the two transients(high to low and low to high) and of the average value of Vx. The second“master” signal (and thus, the remaining fourth required switchingsignal) is generated by the second control circuit (84) for controllingthe switching of the low side switch and is based on the electricalfeedback parameter from the converter input or output in order toprovide the control of the converter's power factor and/or outputvoltage or current. This signal determines the switching frequency,which either is generated explicitly in case of a frequency control orimplicitly, in case of a threshold detection of a self-oscillatingresonant tank circuit.

In other cases, it is beneficial to operate with a defined duty cyclethat is different from 50% in order to enlarge the output voltage orcurrent window that the converter is capable of handling.

For threshold-based resonant converters (such as a self-oscillating LLCconverters), there is no oscillator present in the circuit.Threshold-based switching has a particular advantage with regards to thelinearity of the transfer function when using the converter to cover awide range of input and output operating conditions, such as in an LLCPFC for example, and frequency control is not feasible in such cases dueto extreme variations in the gain that cannot easily be handled.

The approach to the generation of the required voltages for switchingthe high side and low side power switches as explained above may be usedin all of these situations.

The invention may be used in various applications, such as LED driversin general, and in particular front end (isolating) converters forstandalone drivers (indoor and outdoor), particularly miniaturized orflat types, offline drivers for track lighting, emergency lightingdrivers, and miniature, isolating single stage LED drivers. Theconverter may also be used in single-stage separated extra low voltage(SELV) power converters for fixed output voltages and generally inconsumer and office electronics applications such as laptop adapters.

Other variations to the disclosed embodiments can be understood andeffected by those skilled in the art in practicing the claimedinvention, from a study of the drawings, the disclosure, and theappended claims. In the claims, the word “comprising” does not excludeother elements or steps, and the indefinite article “a” or “an” does notexclude a plurality. The mere fact that certain measures are recited inmutually different dependent claims does not indicate that a combinationof these measures cannot be used to advantage. Any reference signs inthe claims should not be construed as limiting the scope.

The invention claimed is:
 1. A half bridge resonant converter,comprising: a pair of DC voltage lines arranged to provide a busvoltage, wherein the pair of DC voltage lines comprises a high voltageline and a low voltage line; a half bridge inverter in series betweenthe high voltage line and the low voltage line, wherein the half bridgeinverter comprises a high side switch and a low side switch, wherein anoutput of the half bridge inverter is defined from a switch node betweenthe high side switch and the low side switch; a resonant circuit coupledto the output of the half bridge inverter; a first control circuit forgenerating a first gate drive signal for controlling switching of thehigh side switch, wherein the first control circuit is arranged tocontrol a duty cycle of the high side switch by increasing an on-time ofthe high side switch if an average switch node voltage is lower than afraction of the bus voltage and by decreasing the on-time of the highside switch if the average switch node voltage is higher than thefraction of the bus voltage; and a second control circuit for generatinga second gate drive signal for controlling switching of the low sideswitch in dependence on an electrical feedback parameter.
 2. The halfbridge resonant converter as claimed in claim 1 wherein the firstcontrol circuit is arranged to turn on the high side switch after thelow side switch is turned off and a dead-time has elapsed.
 3. The halfbridge resonant converter as claimed in claim 1 wherein the secondcontrol circuit is arranged to turn on the low side switch after thehigh side switch is turned off and a dead-time has elapsed.
 4. The halfbridge resonant converter as claimed in claim 1 wherein the secondcontrol circuit is arranged to control at least one of an output powerand a power factor of the half bridge resonant converter by controllingthe low side switch.
 5. The converter as claimed in claim 1, wherein thefirst control circuit comprises: a first end of slope detection circuithaving the high voltage line as an input; a first latch elementtriggered by the end of slope detection circuit and which generates afirst control signal for switching the high side switch to a firststate; and a first signal generator for generating a second controlsignal for switching the high side switch to a second state.
 6. Theconverter as claimed in claim 5, wherein the first signal generator hasa reference input for controlling a duration of the first state.
 7. Theconverter as claimed in claim 1, wherein the second control circuitcomprises: a second end of slope detection circuit having the switchnode as an input; a second latch element triggered by the second end ofslope detection circuit and which generates a third control signal forswitching the low side switch to a first state; and a second signalgenerator for generating a fourth control signal for switching the lowside switch to a second state.
 8. The converter as claimed in claim 7,wherein the second signal generator has a feedback control input forcontrolling a duration of the first state in dependence on theelectrical feedback parameter.
 9. The converter as claimed in claim 1,wherein the electrical feedback parameter comprises a voltage which isdependent on an output current delivered by the converter to a load. 10.The converter as claimed in claim 1, wherein the resonant circuitcomprises an LLC circuit.
 11. The converter as claimed in claim 1,wherein the first circuit and the second control circuits each comprisesan integrated circuit.
 12. An apparatus comprising: the half bridgeresonant converter as claimed in claim 1; and an output load.
 13. Theapparatus as claimed in claim 12 wherein the output load is an LEDarrangement of one or more LEDs.
 14. A conversion method, comprising:operating a half bridge inverter comprising a high side switch and a lowside switch between a DC high voltage line and a DC low voltage lineproviding a bus voltage, using a gate drive signal and providing anoutput from a switch node between the high side switch and the low sideswitch; providing the output of the half bridge inverter to a resonantcircuit; generating a first gate drive signal using a first controlcircuit, for controlling a duty-cycle of the high side switch byincreasing an on-time of the high side switch if an average switch nodevoltage is lower than a fraction of the bus voltage and by decreasingthe on-time of the high side switch if the average switch node voltageis higher than the fraction of the bus voltage; and generating a secondgate drive signal using a second control circuit, for controllingswitching of the low side switch in dependence on an electrical feedbackparameter.
 15. The method as claimed in claim 14, further comprisinggenerating a first supply voltage from a high voltage line and from avoltage at the switch node, and generating a second supply voltage fromthe low voltage line and from the voltage at the switch node.